Low Voltage DC Supply Dimmable Ballast for 1x36W T8 Lamp
Keywords:power
/ARTICLES/2003JUN/A/2003JUN27_POW_AN9.PDF 
Topics Covered
Introduction Schematic Diagram
Functional Description Layout Issues
Preheating & Cathode Heating Output Inductor Design
Control IC Auxilliary Component Selection Bill of Materials
Introduction
It is possible to design an effective dimmable ballast based around the IR2159 that is powered from a low voltage
DC supply instead of the AC line. A nondimmable version based around the IR2156 is also possible using the
same basic configuration as described here. The following example shows a ballast for a single 36W T8 lamp
driven from a 30V DC supply. Lower supply voltages are possible however the IR2159 control IC can require up to
13V supplied to the VCC pin before it will operate, also as the current is high large conductors are needed to keep
losses at an acceptable level (for a 36W ballast operating from a 30V supply the input current is around 1.25A).
Functional Description
The ballast control IC and circuitry is powered from the low voltage DC supply via a simple dropper resistor that
provides the required 15.6V VCC supply from the 30V input. This can easily supply enough current for the ballast
control circuitry dissipating only a small amount of power and therefore no charge pump is needed to sustain the
supply. The circuit configuration consists of a pushpull power switching stage as opposed to the usual halfbridge
in mains supplied ballasts. This simplifies the circuit since a high side driver is not needed so the VB pin of the IC
can be connected directly to the VCC, the VS pin can be connected to 0V and no bootstrap diode or capacitor are
necessary. The output section has a stepup transformer with a split primary which produces a high voltage
switching voltage at the secondary that can be connected to a conventional inductor and capacitor ballast output
stage to the lamp. In this system both power switching MOSFET sources are connected to 0V. In order to obtain
the required current level and phase information a sense resistor must be added from the source of the LO side
MOSFET to 0V. The current will be much larger at this point than in a mains powered ballast circuit and so a lower
resistor value is necessary, in this case 0.1R. Since the step up transformer introduces no significant phase shift
and the waveform is almost identical to a current sense signal in a half bridge ballast circuit, this signal can be
used effectively to detect the phase for dimming control. In this system as well as in nondimming designs this
point can be used to monitor the current and detect faults in the normal way allowing the ballast to shutdown if the
lamp fails to ignite correctly as in half bridge ballast circuits.
It is not necessary for the output section to be isolated from the input section and so one side of the secondary can
be referenced back to the 0V rail. We can therefore connect one side of the lamp to 0V allowing us to utilize the SD
APPLICATIONNOTE AN1038
Low Voltage DC Supply Dimmable Ballast
for 1 x 36W T8 Lamp
By Peter Green
www.irf.com 1
International Rectifier 77777 233 Kansas Street El Segundo CA 90245 USA
2 www.irf.com
AN1038
pin of the control IC to detect lamp removal or an open circuit in the lower filament. During dimming the lamp is
prone to produce striations (dark rings that appear to move along the tube). We can remove these by adding some
DC offset to the lamp voltage via R16 which is connected back to the 30V bus. A snubber consisting of R15 and
C10 is also added to reduce ringing overshoot voltages that occur when each MOSFET switches off. The snubber
will also increase the commutation time at switch off so that soft switching can be achieved using the IR2159
which has a fixed dead time of 1.8uS. The MOSFETs used are type IRF540N which have a Vdss rating of 100V
and Rds(on) of 0.044 at 25:C. The peak drain voltage is 60V plus the transient produced by the leakage induc
tance of the step up transformer when at switch off which is comfortably less than 100V limited by the snubber.
Selection of the output L and C values
Using the lamp parameters supplied by International Rectifiers Ballast Design Assistant software we can calcu
late the preheat, ignition and running frequencies for a bus voltage of 300V (note that version 2 and above will
calculate the frequency for 2% lamp power whereas version 1 will not). The output from the step up transformer
will be a square wave of 300Vpp which is what would be obtained using a half bridge connected to a 300V DC
bus. The output section will be the same as in a half bridge circuit. The DC blocking capacitor must also be
included so that the step up
transformer will not drift into
saturation in one direction if the
primary voltseconds are not
perfectly balanced in each
direction.
In this example the step up
transformer is designed to
operate at 40kHz minimum
frequency where the ballast
will be at maximum output.
The core needs to be larger to
cope with the same through
put power at a lower frequency
so in this case in order to
limit the size to EF25 we have
chosen a 40kHz running fre
quency. By iterating the values
of the L and C in the software
we are easily able find values
that produce the desired run
ning frequency. The values are
L= 1.6mH and C=6.8nF.
Figure 1 The BDA software generates a curve showing the ballast operating points
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AN1038
These points are good because the preheat frequency is more than 5kHz above the ignition frequency which will
prevent the possibility of premature lamp ignition during preheat. The run frequency is as required and is well
below the ignition frequency which will allow a smooth lamp startup sequence.
Preheating and Cathode heating
Current mode preheating may be used in a dimming ballast designed for a 36W T8 lamp as it is able to produce
the correct preheat current which is 0.6A. The required preheat frequency can be obtained from the formula
and
where Vin will be 300V
In a dimming ballast it is also very important that the cathode current is sufficient when the lamp is dimmed.
The Cathode current at minimum can be calculated with the formula
The lamp voltage at 2% is 165Vpk and the frequency is 71kHz therefore the cathode current is 0.5Apk which is
0.35Arms.
A general rule is that the lamp filament (Cathode) resistance over the range of dimming levels should be between
3 and 5.5 times the resistance when cold. For a T8 36W lamp the cold resistance will be around 3. The resis
tance of the cathode at maximum brightness is not critical as the arc current flowing in the lamp will serve to keep
the temperature at a sufficient level. At minimum output the cathode voltage will be around 3Vrms so the resis
tance will be 9 which is 3 times the cold resistance. This will not be the case for many other types of lamp and
consequently voltage mode preheating would be needed.
Control IC auxiliary component selection
The quickest and easiest method for doing this in each case is to use version 2 (or higher) of the International
Rectifier Ballast Design Assistant software which can be downloaded from IR's website at www.irf.com. The BDA
supports both the IR2156 and the IR2159 as well as the IR21571 which could also be used in a non dimming low
voltage design. It can calculate the values of all external resistors and capacitors using the procedures described
below.*
(Please note that version 1 of the BDA is not suitable for using in this application)
ph
ph
ph
CV
i
f
2
=
++

=
22
C
ILVV
V
phinin
ph
CfVI lampcath %2%)2(
2=
4 www.irf.com
AN1038
IR2156 based system :
For non dimming design based around an IR2156 the selection of components is straightforward. CT should be
selected to provide a dead time of approximately 1.85S which is the same as the fixed dead time of the IR2159.
This can be calculated from the formula :
(Farads)
The value obtained is 1.2nF, in practise 1nF would be acceptable. The next step is to determine the value of RT
which can be calculated from the formula :
() For Frun = 40kHz the value of RT is 22K. *
The value of RPH can be calculated from the formula :
() *
The preheat time is determined by the value of CPH calculated from the formula :
(Farads) 0.33uF is the value typically used to give a 1S preheat time.
In this circuit RCS is used to shut down the circuit in a fault condition. The shutdown threshold is 1.3V therefore
the value can be calculated from the modified formula :
()
.
where Np is the transformer primary turns (center to one side) and Ns is the secondary turns.
In this case, taking a value of 2A for the ignition current the value is 0.13 so we scale this down to the nearest
preferred value 0.1.
The VDC pin can be used by connecting it via a resistor to the DC bus so that if the supply voltage falls the output
frequency will increase preventing the possibility of hard switching occurring which would cause overheating and
possible failure of the MOSFETs. A value should be chosen that will start to take effect at around 25V in a system
designed to run at 30V in this case 150K is recommended.
1475
T
T
D
C =
2892
02.1
1

=
runT
T
fC
R



=
2892
02.1
1
2892
02.1
1
PHT
T
PHT
T
PH
fC
R
fC
R
R
6
10385.0 
W= PHPH TC
SIGN
P
CS
NI
N
R
3.1
=
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AN1038
IR2159 based system :
In a dimming system the selection of external components is more critical and the procedure more complicated.
The values of Fph, Fign, F(100%) and F(2%) can be calculated by hand using the procedure described in the
Lighting Ballast Control IC Designer's Manual 2001* based on known values of lamp voltage and power. However
the BDA software is able to do this far more quickly using lamp parameters from its own database.
* Please note that there is an error on p.213 of the Lighting Ballast Control IC Designer's Manual (2001) the
formulae for f(100%) and f% should be as stated on p250 and p.251
The following procedure can be used to determine the values for the IR2159 external resistors. Firstly Fmin must
be set which limits the minimum frequency that the oscillator will run at. This must be lower than F(100%) or Fign,
whichever is lower.
()
In this case we can select Fmin to be 40kHz this will limit the maximum ballast power. This gives a value of 36K.
The next step is to determine RCS from the formula
()
For an ignition current of 2A the value will be 0.16 which can be scaled down to 0.15 or 0.1. The over current
shutdown will obviously be more sensitive for 0.15 so if no problems of false tripping are experienced this would
give better protection.
If using the BDA software to calculate the resistor via the Program IC function it should be noted that the step up
function of the transformer will not be taken into account. Therefore the value given for RCS will need to be
multiplied by Np/Ns to provide the correct value to use in a low voltage system. This is unlikely to result in a
preferred E24 resistor value. The easiest way around this is to choose the nearest preferred value and multiply Ns/
Np. Make a note of the result and then adjust the ignition current in the software and recalculate until the RCS is the
same as the value required. Notice that the values of Riph, Rmax and Rmin will change as RCS changes. In this
example the value of 0.15 is multiplied by 125/25 giving 0.75. The ignition current is changed from 1.8A to 2.2A
and the resistor values recalculated to give RCS of 0.75.
( )
( ) 14
106
10210000
10100001025


W
W
=
MIN
MIN
FMIN
f
f
R
SIGN
P
CS
NI
N
R
6.1
=
6 www.irf.com
AN1038
The next step is to calculate Riph from the following formula
()
For a preheat current of 0.6Arms which is correct for this lamp the result is 22K.
In order to program the MIN and MAX settings of the dimmer interface, the phase of the output current stage at
minimum and maximum lamp power must be calculated. This is a very complicated calculation requiring the lamp
voltage and power to be known at minimum and maximum dim settings. The following method avoids the need for
this by assuming that the phase will be very close to 90: at minimum brightness and using this value to calculate
Rmin from the formula
() This give the result 27K
The BDA software can calculate the value of Rmax by first calculating the phase shift based on its database lamp
data parameters, however to get a rough estimate of what Rmax should be we can use an approximation of the
phase shift. We know from graph of Fig 1 that the frequency at 100% power is below resonance and so the phase
shift must be between 45: and zero. To obtain a starting point we can estimate a phase shift of 30: at maximum
brightness and use the formula
() This give the result 18K
Having determined the values the circuit can then be assembled and tested. By measuring the input current to the
ballast it is possible to know whether the maximum power is correct (i.e. the current is 1.25A) and if not to change
the value of Rmax accordingly. It is possible to observe the lamp brightness at minimum and adjust the value of
Rmin to obtain the desired result. It is a good idea to build the board fitting multi turn trimmers to R4 (Rmax) and R7
(Riph) to allow fine tuning of the values to give best possible results. (This can be done for Rt and Rph in the non
dimming IR2156 based version.)
* If necessary values can be altered slightly to achieve fine adjustment of the frequencies to obtain the optimum
preheat level and lamp running power.
P
SPHCSFMIN
IPH
N
NIRR
R
2
=
=
45
1
4
%2FMIN
MIN
R
R

=
45
14
86.0
%100
FMINMIN
MINFMIN
MAX
RR
RR
R
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AN1038
Schematic Diagram
Layout Issues
When laying out the PCB for this type of ballast it is important to allow the high current carrying tracks to be as wide
and as short as possible. The control IC COM should be a star point connected to the COM end of R12. The IC
COM should be connected directly to this point via a very short track. The negative side of C2 should also be
connected as close to this point as possible and the positive side should be connected as close to the center tap
of T1 as possible. The C9 decoupling capacitor should be connected directly between VCC and COM and C3, C4,
C5, R4, R5, R6, R7, C6, C7, C8 should all be connected back to the star point. Tracks around the IC should be
kept short as far as possible except the gate drives and VS and VB which can be a little longer if necessary. It is
also important to keep tracks that are carrying high switching currents away from sensitive components around
the IC as much as possible.
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
IR2159
VDC
CPH
DIM
MAX
MIN
FMIN
IPH
LO
COM
VCC
VB
VS
HO
SD
CS
VCO
1uF
C1 C2
220uF
L1
Control Input
0 to 5Vdc
(w.r.t. 0V)
+30VDC
0V
36W
T8
LAMP
R1
24K
R2
5K6
R3 10K
R4 12K
R5 27K
R6 36K
R7 27K
R8
1K
0.5W
R9
2M2
R10
680K
R11
1K
R12
0R15
R13
18R
R14
18R
Q1
IRF540N
Q2
IRF540N
R15
1K5
C3100nF
C4 10nF
C5 330nF
C6
470nF
C7
470pF
C8 C9
100uF
25V
100nF
NOTE : All capacitors are rated at 50V DC unless otherwise
stated.
C10
100nF
400V
1W
C11
100nF 400V
L2
1.6mH
C12
6n8
1500V
50V 50V
25+25 : 125
R16
22K
1W
IC1
F1
2A
T1
8 www.irf.com
AN1038
StepUp Transformer Design
Voltage across the primary winding (from drain to drain)
The above oscilloscope traces show the voltage at the drain of each of the switching MOSFETS. The drain voltage
rises to 60V when the MOSFET switches off. This is because the primary winding is center tapped and the center
point is connected to the 30V DC bus. When one MOSFET is switched on the voltage between the center point
and the drain is 30V therefore the voltage across the entire primary winding will swing from 60V in one direction to
60V in the other direction the result being 120Vpp.
In this design we have chosen the turns ratio of the transformer to give 300Vpp at the secondary which can be fed
into the ballast resonant output circuit. The turns ratio required can be determined as follows :
300 / 120 = 2.5
2 x 2.5 = 5
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AN1038
Therefore the turns ratio will be 1 + 1 : 5.
The core size should be selected for a throughput power of 36W at 40kHz. We have used an EF25 (E25/13/7)
core of 3C85 or N27 material which ungapped has an Al value of 1900nH and an effective area Ae of 52mm2.
Primary VoltSeconds = 60V x 12.5uS = 750VuS
We have chosen 25 + 25 : 125 turns.
This gives a primary inductance of 502 x 1900nH = 4.75mH.
Therefore the magnetizing current will be 750 x 106 / 4.75 x 103 = 0.16A.
The peak flux will be
= 50 x 1900 x 109 x 0.16 / 52 x 106 = 0.29T (2900 Gauss).
This shows that the core is being pushed close to saturation in each direction but will not saturate at high
temperature (see manufacturers BH curve for the Ferrite material).
The winding wire sizes should be chosen such that they fill the winding area. The primary should have approxi
mately twice the diameter of the secondary as the primary RMS current will be 1.25A and the secondary RMS
current will be 0.25A.
e
PKLP
A
IAN
10 www.irf.com
AN1038
Voltage across the secondary winding
Output Inductor Design
The BDA software will design the output inductor if required. It will suggest a wire diameter for a single strand,
however a multi stranded wire that has an equivalent total cross sectional are will produce lower copper losses.
Alternatively the following procedure may be used :
1. Select the winding wire
In a dimming design because the frequency goes as high as 70kHz it is necessary to use multi stranded wire in
order to keep losses due to the skin effect to a minimum. If single stranded wire is used the inductor will run at an
increased temperature when the lamp output is low.
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AN1038
Consider the RMS running current of the lamp which can be easily estimated by dividing the maximum lamp
power by the RMS lamp voltage. The RMS lamp voltage can be approximated by dividing the peak lamp voltage by
v2 in this case 100V giving 0.36A. A current density 3A/mm2 can be used to calculate to minimum cross sectional
area of conductor that will be required. In this case the result is 0.12mm2.
The skin effect must now be considered. For Copper conductors the penetration depth at a given frequency can
be calculated by the formula
(mm)
Using the maximum frequency of 70kHz this gives the result 0.24mm therefore the strands should be less than
0.24mm diameter.
The conductor area for a wire of 0.24mm is
This gives the result 0.073mm2.
A practical solution would be to use 4 strands wire that has a diameter much smaller than 0.24mm. The area for
each strand would have to be 0.03mm2 this equates to AWG 32 which has an area of 0.046mm2 including the
insulation.
2. Select the core size
The BDA uses an iterative process which tries a range of core and gap sizes finally selecting the smallest size
that can contain the winding wire without saturating during lamp ignition. This is extremely important because if the
core does saturate the resulting current pulse will be detected at the CS pin of the IC causing the ballast to shut
down. A common design error is to fail to allow for a hot restrike condition (i.e. when the ballast has been running
and is switched off and back on again) where the Ferrite core is already at increased temperature and the satura
tion point of the material is reduced resulting in saturation at a lower current.
To follow the procedure of the BDA by hand is time consuming and therefore it is easier to pick an option based
on experience. For a 36W ballast a reasonable starting point would be to design an inductor based on an EF25
(E25/13/7) core with a standard gap size of 1mm made of standard power grade Ferrite (type 3C85 or N27).
For this the Al value is 63nH and Ae is 52mm2. The inductance required is 1.6mH therefore
The number of turns required is 159.
f
65
=
2
4D
A =
LA
L
N =
12 www.irf.com
AN1038
The maximum ignition current is 2A so the peak flux density will be
Which gives the result 0.39T (3900 Gauss). By looking at the manufacturers curve of B against H we can see that
the material will saturate at around 0.42T at 25:C and 0.35T at 100:C. When the ballast is cold there is no possi
bility of saturation at ignition and during a hot restrike situation the core is unlikely to be as hot as 100:C. Therefore
this solution is acceptable as in reality the ignition voltage of the lamp will be somewhat less than 2A if the lamp is
correctly preheated. The inductor should be built and tested under worst case conditions to ensure that the lamp
will strike. If there are problems then a larger gap or larger core will be required.
The available winding area in an EF25 bobbin is 56mm2. The winding area required is
0.046 x 4 x 159 = 29.3mm2
Allowing for gaps there will be plenty of room. It is always an advantage to increase to wire size or better still add
more strands as much as possible to minimize copper losses when the lamp is running. The BDA does this
automatically.
Bill of Materials
The following component values have been selected for a 36W T8 lamp only.
e
PKL
MAX
A
INA
B =
Item
#
Qty Manufacturer Part
Number
Description Reference
1 2 I.R. IRF540 Power MOSFET Q1,2
2 1 I.R. IR2159 Ballast Control I.C. IC1
3 1 Fuse 2A F1
4 1 Capacitor 1uF 50V 105:C
Radial Electrolytic
C1
5 1 Capacitor 220uF 50V 105:C
Radial Electrolytic
C2
6 2 Capacitor 100nF 50V C3,8
7 1 Capacitor 10nF 50V C4
8 1 Capacitor 330nF 50V C5
9 1 Capacitor 470nF 50V C6
10 1 Capacitor 470pF 50V C7
11 1 Capacitor 100uF 25V 125:C
Radial Electrolytic
C9
www.irf.com 13
AN1038
WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245 Tel: (310) 2527105
http://www.irf.com/ Data and specifications subject to change without notice. 4/3/2002
y
12 2 Capacitor 100nF 400V
Polyester
C10,11
13 1 Capacitor 6.8nF 1500V
Polypropylene
C12
14 1 Resistor 24K 0.25W R1
15 1 Resistor 5K6 0.25W R2
16 1 Resistor 10K 0.25W R3
17 1 Resistor 12K 0.25W R4
18 2 Resistor 27K 0.25W R5,7
19 1 Resistor 36K 0.25W R6
20 1 Resistor 1K 0.5W R8
21 1 Resistor 2M2 0.25W R9
22 1 Resistor 680K 0.25W R10
23 1 Resistor 1K 0.25W R11
24 1 Resistor 0R15 0.25W R12
25 2 Resistor 18R 0.25W R13,14
26 1 Resistor 1K5 0.25W R15
27 1 Resistor 22K 1W R16
28 1 Filter Inductor L1
29 1 Inductor 1.6mH EF25 L2
30 1 Transformer 1+1:5 EF25 T1
Item
#
Qty Manufacturer Part
Number
Description Reference
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