Design of power factor correction circuit using Greenline compact power factor controller MC33260

) Semiconductor Components Industries, LLC, 1999 February, 2000 - Rev. 0 1 Publication Order Number: AND8016/D AND8016/D Design of Power Factor Correction Circuit Using GreenlineTM Compact Power Factor Controller MC33260 Prepared by Ming Hian Chew ON Semiconductor Analog Applications Engineering Introduction The MC33260 is an active power factor controller that functions as a boost pre-converter which, meeting international standard requirement in electronic ballast and off-line power supply application. MC33260 is designed to drive a free running frequency discontinuous mode, it can also be synchronized and in any case, it features very effective protections that ensure a safe and reliable operation. This circuit is also optimized to offer extremely compact and cost effective PFC solutions. It does not entail the need of auxiliary winding for zero current detection hence a simple coil can be used instead of a transformer if the MC33260 Vcc is drawn from the load (please refer to page 19 of the data sheet). While it requires a minimum number of external components, the MC33260 can control the follower boost operation that is an innovative mode allowing a drastic size reduction of both the inductor and the power switch. Ultimately, the solution system cost is significantly lowered. Also able to function in a traditional way (constant output voltage regulation level), any intermediary solutions can be easily implemented. This flexibility makes it ideal to optimally cope with a wide range of applications. This application note will discuss on the design of power factor correction circuit with MC33260 with traditional boost constant output voltage regulation level operation and follower boost variable output voltage regulation level operation. For derivation of the design equations related to the IC please refer to MC33260 data sheet. Figure 1. Application Schematic of MC33260 1 2 3 4 8 7 6 5 MC33260 C5 + D5 Q1R5 R4R3 R2 C1 D1 D3 D2 D4 + C4 R6 D5 D7 R7 R1 C2 C3 L1 C6 PFC Techniques Many PFC techniques have been proposed, boost topology, which can operate in continuous and discontinuous mode, is the most popular. Typically, continuous mode is more favorable for high power application for having lower peak current. On the other hand, for less than 500W application, discontinuous mode offers smaller inductor size, minimal parts count and lowest cost. This paper will discuss design of PFC with MC33260, which operates in critical conduction mode. Discontinuous Conduction Mode Operation Critical conduction mode operation presents two major advantages in PFC application. For critical conduction mode, the inductor current must fall to zero before start the next cycle. This operation results in higher efficiency and http://onsemi.com APPLICATION NOTE AND8016/D http://onsemi.com 2 eliminates boost rectifier reverse recovery loss as MOSFET cannot turn-on until the inductor current reaches zero. Secondly, since there are no dead-time gaps between cycles, the ac line current is continuous thus limiting the peak switch to twice the average input current. The converter works right on critical conduction mode, which results in variable frequency operation. Inductor Waveform V L + di dt (1) Equation (1) is the center of the operation of PFC boost converter where V=Vin(t), the instantaneous voltage across the inductor. Assuming the inductance and the on-time over each line half-cycle are constant, di is actually the peak current, ILpk, this is because the inductor always begins charging at zero current. Figure 2. Inductor Waveform Vinpk ILpk Iinpk Vin(t) IL(t) Iin(t) ON OFF MOSFET Design Criteria The basic design specification concerns the following: 7 Mains Voltage Range: Vac(LL) - Vac(HL) 7 Regulated DC Output Voltage: Vo 7 Rated Output Power: Po 7 Expected Efficiency, h PFC Power Section Design Instantaneous Input Voltage, Vin(t) Peak Input Voltage, Vinpk Both Vin(t) and Vinpk are related by below equation V in (t) +V inpk sin(t) (2) V inpk + 2 Vinrms (3)where Instantaneous Input Current, Iin(t) Peak Input Current, Iinpk, Both Iin(t) and Iinpk are related by below equation I in (t) +I inpk sin(t), (4) I inpk + 2 Iinrms (5)where Input power of the PFC circuit, Pin can be expressed in following equation, by substituting equation (3) and (5). P in +V inrms I inrms + V inpk 2 @ I inpk 2 + V inpk I inpk 2 (6) The output power, Po is given by: Po +VoIo +P in (7) PFCcircuitefficiencyisneededinthedesignequation,for low line operation, it is typically set at 92% while 95% for high line operation. Substituting equation (6) into equation (7), Po +P in + V inpk I inpk 2 (8) Express the above equation in term of Iinpk, I inpk + 2Po V inpk + 2 Po V inrms (9) The average input current is equal to average inductor current, IL(avg), I L(avg) +I in (10) It has been understood that peak inductor current, ILpk is exactlytwicetheaverageinductorcurrent,IL(avg)forcritical conduction operation. I Lpk +2I L(avg) + 2 2 Po V inrms (11) Since ILpk is maximum at minimum required ac line voltage, Vac(LL), I Lpk + 2 2 Po V ac(LL) (12) Switching Time In theory, the on-time, t(on) is constant. In practice, t(on) tends to increase at the ac line zero crossings due to the chargeonoutputcapacitorCout.LetVac=Vac(LL)forinitial t(on) and t(off) calculations. On-time By solving inductor equation (1), on-time required to charge the inductor to the correct peak current is: t (on) +I Lpk L P Vinpk (13) Substituting equation (3) and (12) into equation (13), results in: t (on) + 2 2 Po V ac(LL) @ L P 2 Vac(LL) + 2PoL P V2 ac(LL) (14) Off-time Theinstantaneousswitchoff-time varies with the line and load conditions, as well as with the instantaneous line voltage. Off-time is analyzed by solving equation (1) for the inductor discharging where the voltage across the inductor is Vo minus Vin. AND8016/D http://onsemi.com 3 t (off) + I Lpk L P Vo *V inpk sin(t) (15) Multiplying nominator and denominator with Vinpksinw(t) results in: t (off) + (16) I Lpk L P V inpk sin(t) Vo *V inpk sin(t) V inpk sin(t) + Vo 2 Vinpk sin() *1 t (on) where wt = The off-time, t(off) is greatest at the peak of the ac line voltage and approaches zero at the ac line zero crossings. Theta () represents the angle of the ac line voltage. The off-time is at a minimum at ac line crossings. This equation is used to calculate t(off) as Theta approaches zero. t (off)min + I Lpk L P Vo , +00 (17) Switching Frequency f + 1 t (on) )t (off) (18) Switching frequency changes with the steady state line and load operating conditions along with the instantaneous input line voltage. Typically, the PFC converter is designed to operate above the audible range after accommodating all circuit and component tolerances. 25kHz is a good first approximation. Higher frequency operation that can significantly reduce the inductor size without negatively impacting efficiency or cost should also be evaluated. The minimum switching frequency occurs at the peak of theaclinevoltage.Astheaclinevoltagetraversesfrompeak to zero, t(off) approaches zero producing an increase in switching frequency. Inductor Value Maximum on-time needs to be programmed into the PFC controllertimingcircuit.Botht(on)maxandt(off)maxwillbe individually calculated and added together to obtain the maximumconversionperiod,ttotal.Thisisrequiredtoobtain the inductor value. t (on)max + 2PoL P V 2 ac(LL) (19) t (off)max + I Lpk L P Vo *V inpk , () [900 (20) The exact inductor value can be determined by solving equation (21) by substituting equation (19) and (20) at the selected minimum operating frequency. t total +t (on)max )t (off)max (21) Equation (21) becomes t total + 2 PoL P Vo V 2 ac(LL) Vo 2 * V ac(LL) (22) By rearranging in term of Lp, Lp + t total Vo 2 * V ac(LL) V 2 ac(LL) 2 VoPo (23) Equation (23) can be rewritten by substituting rearranged equation (12) in term of 2Po. Lp + 2 t total Vo 2 * V ac(LL) Vac(LL) VoI Lpk (24) Let the switching cycle t = 405s for universal input (85 to 265Vac)operationand205sforfixedinput(92to138Vac,or 184 to 276Vac) operation. Inductor Design Summary The required energy storage of the boost inductor is: W L + 1 2 L P I 2 Lpk (25) The number of turns required for a selected core size and material is: N P + L P I Lpk 106 BmaxAe (26) where Bmax is in Teslas and Ae is in square millimeters (mm2) The required air gap to achieve the correct inductance and storage is expressed by: lgap +4p10 *7 N 2 p Ae L P mm (27) Design of Auxiliary Winding MC33260 does not entail an auxiliary winding for zero current detection. Hence if DC voltage can be tapped from the SMPS or electronic ballast connected to the output of PFC, this step can be skipped. Then an inductor is what it needs. The auxiliary winding will have the low frequency (100-120Hz) variation on its peak-rectified voltage. The auxiliary winding capacitor must be large to minimize the droop in the Vcc of the controller. The auxiliary turn number can be approximated with below equation: Naux +VauxN P VL + VauxN P Vo *V ac(LL) (28) Selection of Output Capacitor The choice of output capacitance value is dictated by the required hold-up time, thold or the acceptable output ripple AND8016/D http://onsemi.com 4 voltage,Voripforagivenapplication.Asaruleofthumb,can start with 15F/watt. Selection of Semiconductors Maximum currents and voltages must first be determined for over all operating conditions to select the MOSFET and boost rectifier. As a rule of thumb, derate all semiconductors to about 75-80% of their maximum ratings. This implying the need of devices with at least 500V breakdown voltage. BipolartransistorsareanacceptablealternativetoMOSFET if the switching frequency is maintained fairly low. High voltage diodes with recovery times of 200ns, or less should be used for the boost rectifier. One series of the popular devices is the MURXXX Ultrafast Rectifier Series from ON Semiconductor. Maximum power MOSFET conduction losses. P (on)max [1 6 R ds(on) I 2 Lpk 1 * 1.2 V ac(LL) Vo (29) Designing the Oscillator Circuit For traditional boost operation, CT is chosen with below equation: C T w2 Kosc L P P in V2 o V 2 ac(LL) R 2 o *Cint (30) Design of Regulation and Overvoltage Protection Circuit The output voltage regulation level can be adjusted by Ro, Ro [ Vo 200 5A (31) Designing the Current Sense Circuit The inductor current is converted into a voltage by insertinga ground referenced resistor, RCS in series with the input diode bridge. Therefore a negative voltage proportional to the inductor current is built. The current sense resistor losses, PRcs: P Rcs +1 6 R CS I 2 Lpk (32) Overcurrent protection resistor, ROCP can be determined with below equation: R OCP + R CS I Lpk I OCP (33) Current Limiting With Boost Topology Power Factor Correction Circuit Unlike buck and flyback circuits, because there is no seriesswitchbetweeninputandoutputintheboosttopology, high current occurring with the start-up inrush current surge charging the bulk capacitor and fault load conditions cannot be limited or controlled without additional circuitry. The MC33260 Zero Current Detection uses the current sensing information to prevent any power switch turn on as long as some current flows through the inductor. Then, during start-up, the power MOSFET is not allowed to turn on while in-rush current flows. Then there is no risk to have the power switch destroyed at start-up because of the in-rush current. In the same way, in an overload case, the power MOSFET is kept off as long as there is a direct output capacitor charge current, i.e., when the input voltage is higher than the output voltage.Consequently, overload working is fully safe for the power MOSFET. This is one of the major advantages compared to MC33262 and competition. Current Limiting for Start-up Inrush Initially Vo is zero, when the converter is turned on, the bulk capacitor will charge resonantly to twice Vin. The voltage can be as high as 750V if Vin happens to be at the peak high-line 265V condition (375V). The peak resonant charging current through the inductor will be many times greater than normal full load current. the inductor must be designed to be much larger and more expensive to avoid saturation. The boost shunt switch cannot do anything to prevent this and could be worse if turned on during start-up. The inrush current and voltage overshoot during the start-up phase is intolerable. A fuse is not suitable, as it will blow each time the supply is turned on. There are several methods that may be used to solve the start-up problem: 1. Start-up Bypass Rectifier This is implemented by adding an additional rectifier bypassingthe boost inductor. The bypass rectifier will divert the start-up inrush current away from the boost inductor as shown in Figure 3. The bulk capacitor charges through Dbypass to the peak AC line voltage without resonant overshootandwithoutexcessiveinductorcurrent.Dbypassis reverse-biased under normal operating conditions. If load overcurrent pulls down Vo, Dbypass conducts, but this is probably preferable to having the high current flowing through boost inductor. Figure 3. Rectifier bypass of start-up inrush current VAC VOUT Dbypass +PFC IC 2. External Inrush Current Limiting Circuit For low power system, a thermistor in series with the pre-converter input will limit the inrush current. Concern is the thermistor may not respond fast enough to provide protection after a line dropout of a few cycles. AND8016/D http://onsemi.com 5 A series input resistor shunted by a Triac or SCR is a more efficient approach. A control circuit is necessary. This method can function on a cycle-by-cycle basis for protection after a dropout. Load Overcurrent Limiting If an overcurrent condition occurs and exceeds the boost converter power limit established by the control circuit, Vo will eventually be dragged down below the peak value of the AC line voltage. If this happens, current will rise rapidly and without limit through the series inductor and rectifier. This may result in saturation of the inductor and components will fail. The control circuit holds off the shunt switch, since the current limit function is activated. It cannot help to turn the switch ON - the inductor current will rise even more rapidly and switch failure will occur. Typically, a power factor correction circuit is connected to another systems like switched mode power supply or electronic ballast. These downstream converters typically will have current limiting capability, eliminating concern about load faults. However, a downstream converter or the bulk capacitor might fail. Hence there is a possibility of a short circuit at the load. If it is considered necessary to limit the current to a safe value in the event of a downstream fault, some means external to the boost converter must be provided. Design Example I - Traditional Boost Constant Output Voltage Regulation Level Operation Power Factor Correction The basic design specification concerns the following: 7 Mains Voltage Range: Vac(LL) - Vac(HL) = 85 - 265Vac 7 Regulated DC Output Voltage: Vo = 400Vdc 7 Rated Output Power: Po = 80W 7 Expected Efficiency, h > 90% A. The input power, Pin is given by P in +Po + 80 0.92 +86.96W B. Input diode current is maximum at Vinrms = Vac(LL) I inpk + 2 Po V ac(LL) + 2 80 0.92 85 +1.447A C. Inductor design 1. Inductor peak current: I Lpk +2I inpk +2 1.447 +2.894A 2. Inductor value: Lp + 2 t total Vo 2 * V ac(LL) Vac(LL) VoI Lpk + 2 40 10*6 400 2 * 85 85 400 2.894 +1.162mH Let the switching cycle t = 40ms for universal input (85 to 265Vac) operation. 3. The number of turns required for a selected core size and material is: N P + L P I Lpk 106 BmaxAe +1.162 10*3 2.894 10*6 0.3 60 +186.8 turns [187 turns Using EPCOS E 30/15/7, Bmax =0.3T and Ae = 60mm2. 4. The required air gap to achieve the correct inductance and storage is: lgap +4p10 *7 N 2 pAe L P +4p 10*7 1872 60 10*6 1.162 10*3 +2.269mm 5. Design of Auxiliary Winding Naux + VauxN P Vo *V ac(LL) + 14 187 (400 *265) +19.4 turns [20 turns Round up to 20 turns to make sure enough voltage at the auxiliary winding. D. To determine the output capacitor As rule of thumb, for 80W output, start with 100mF, 450V capacitor. E. Calculation of MOSFET conduction losses A 8A, 500V MOSFET, MTP8N50E is chosen. The on resistance, Rds(on)[[email protected] Therefore, maximum power MOSFET conduction losses is: P (on)max [1 6 R ds(on) I 2 Lpk 1 * 1.2 V ac(LL) Vo +1 6 1.75 2.8942 1 *1.2 85 400 +1.82W F. Design of regulation and overvoltage protection circuit The output voltage regulation level can be adjusted by Ro, Ro [ Vo 200 5A + 400 200 5A +2M Use two 1MW resistors in series. G. Designing the oscillator circuit For traditional boost operation, CT is chosen with below equation: C T w2 Kosc L P P in V2 o V2 ac(LL) R2 o *Cint + 2 6400 1.162mH 86.96 4002 852 2M2 *15pF +7.16nF AND8016/D http://onsemi.com 6 Use 10nF capacitor. Figure 4. Theoretical Vo versus Vac with CT = 10nF Vo/(V) 85 115 175 85 145 Vac (V) 145 175 115 445 205 235 205 235 265 295 325 355 385 415 265100 1130 190160 220 250 280 Full Load Half Load Vacpeak H. Design of the current sense circuit Choose Rcs = 0.68W 1. So the current sense resistor losses, PRcs: P Rcs +1 6 R CS I 2 Lpk +1 6 1 2.894 2 +0.949W Therefore the power rating of RCS is chosen to be 2W. 2. Overcurrent protection resistor, ROCP can be determined with below equation: R OCP + R CS I Lpk I OCP +0.68 2.894 205 5A +9600 Use 10000W resistor. This provide current limit at 3.01A versus calculated value of ILpk = 2.894A. 80W, Universal Input, Traditional Boost Constant Output Voltage Level Regulation Operation Power Factor Correction Circuit Part List Index Value Comment Index Value Comment C1 [email protected] Filtering Capacitor R6 [email protected] Aux Winding Resistor C2 680nF Pin 2 Vcontrol Capacitor R7 [email protected] Start-up Resistor C3 10nF Pin 3 Oscillator Capacitor R8 1N5406 Input Diode C4 [email protected] Aux Capacitor, E-Cap D1 1N5406 Input Diode C5 [email protected] Output Capacitor, E-Cap D2 1N5406 Input Diode C6 [email protected] Feedback Filtering Capacitor D3 1N5406 Input Diode R1 [email protected] Current Sense Resistor D4 1N4937 Aux Winding Diode R2 [email protected] OCP Sensing Resistor D5 MUR460 Boost Diode R3 [email protected] Feedback Resistor D6 1N5245 Aux 15V Zener Diode R4 [email protected] Feedback Resistor D7 MTP8N50E Power MOSFET R5 [email protected] Gate Resistor Q1 1.162mH Inductor * E 30/15/7, N67 Material from EPCOS Primary - 187 turns of # 23 AWG, Secondary - 19 turns of # 23 AWG. Gap length 2.269mm total for a primary inductance LP of 1.162mH. AND8016/D http://onsemi.com 7 Figure 5. 80W universal input, traditional boost constant output voltage regulation level operation power factor correction circuit 1 2 3 4 8 7 6 5 MC33260 C5 + D5 Q1R5 R4R3 R2 C1 D1 D3 D2 D4 + C4 R6 D5 D7 R7 R1 C2 C3 L1 C6 Design Table for Universal Input, Traditional Boost Constant Output Voltage Regulation Level Operation Power Factor Correction Po 25 50 75 100 125 150 200 (Watts) LP 3.720 1.860 1.240 0.930 0.744 0.620 0.465 (mH) Co 33 68 100 100 150 150 220 (5F) RCS 2 1 0.68 0.5 0.39 0.33 0.25 W ROCP 10000 10000 10000 9100 9100 9100 9100 W Cin 0.22 0.63 0.63 1.0 1.0 1.0 1.0 (5F) CT 10 10 10 10 10 10 10 (nF) Q MTP4N50E MTP8N50E MTW14N50E Dout MUR160 MUR460 Din 1N4007 1N5406 Design Example II - Follower Boost Variable Output Voltage Regulation Level Operation Power Factor Correction The basic design specification concerns the following: 7 Mains Voltage Range: Vac(LL) - Vac(HL) = 85 - 265Vac 7 Maximum Regulated DC Output Voltage: Vo = 400Vdc 7 Minimum Regulated DC Output Voltage: Vomin = 140Vdc 7 Rated Output Power: Po = 80W 7 Expected Efficiency, h > 90% A. The input power, Pin is given by P in +Po + 80 0.92 +86.96W B. Input diode current is maximum at Vinrms = Vac(LL) I inpk + 2 Po V ac(LL) + 2 80 0.92 85 +1.447A C. Inductor design 1. Inductor peak current: I Lpk +2I inpk +2 1.447 +2.894A 2. Inductor value, for follower boost operation, Vo = Vomin: Lp + 2 t total Vomin 2 * V ac(LL) V omin I Lpk + 2 40 10*6 140 2 * 85 85 140 2.894 +0.235 5H Let the switching cycle t = 405s for universal input (85 to 265Vac) operation. 3. The number of turns required for a selected core size and material is: AND8016/D http://onsemi.com 8 N P + L P I Lpk 106 BmaxAe + 0.235 10*3 2.894 106 0.3 32.1 +70.6 turns [71 turns Using EPCOS E 20/10/6, N67 material, Bmax =0.3T and Ae = 32.1mm2. 4. The required air gap to achieve the correct inductance and storage is: lgap +4p10 *7 N 2 pAe L P +4p 10*7 712 32.1 10*6 0.235 10*3 +0.856mm 5. Design of Auxiliary Winding Naux + VauxN P Vo *V ac(LL) + 14 71 (400 *265) +7.4 turns [8 turns Round up to 8 turns to make sure enough voltage at the auxiliary winding. D. To determine the output capacitor As rule of thumb, for 80W output, start with 1005F, 450V capacitor. E. Calculation of MOSFET conduction losses A 4A, 500V MOSFET, MTP4N50E is chosen. The on resistance, Rds(on)[[email protected] Therefore, maximum power MOSFET conduction losses is: P (on)max [1 6 R ds(on) I 2 Lpk 1 * 1.2 V ac(LL) V omin +1 6 1.75 2.8942 1 *1.2 85 140 +0.66W F. Design of regulation and overvoltage protection circuit The output voltage regulation level can be adjusted by Ro, Ro [ Vo 200 5A + 400 200 5A +2M Use two 1MW resistors in series. G. Designing the Oscillator Circuit For follower boost operation, CT is chosen with below equation: C T w2 Kosc L P P in V2 o V2 ac(LL) R2 o *Cint + 2 6400 0.234mH 86.96 1402 852 2M2 *15pF +162pF Use 150pF capacitor. Figure 6. Theoretical Vo versus Vac with CT = 150pF Vo/(V) 85 115 175 85 145 Vac (V) 145 175 115 445 205 235 205 235 265 295 325 355 385 415 265100 1130 190160 220 250 280 Full Load Half Load Vacpeak H. Design of the Current Sense Circuit Choose Rcs = 0.68W 1. So the current sense resistor losses, PRcs: P Rcs +1 6 R CS I 2 Lpk +1 6 0.68 2.8942 +0.949W 2. Overcurrent protection resistor, ROCP can be determined with below equation: R OCP + R CS I Lpk I OCP +0.68 2.894 205 5A +9600 Use 10000W resistor. This provide current limit at 3.01A versus calculated value of ILpk = 2.894A. AND8016/D http://onsemi.com 9 80W, Universal Input, Follower Boost Variable Output Voltage Regulation Level Operation Power Factor Correction Circuit Part List Index Value Comment Index Value Comment C1 [email protected] Filtering Capacitor R6 [email protected] Aux Winding Resistor C2 680nF Pin 2 Vcontrol Capacitor R7 [email protected] Start-up Resistor C3 150pF Pin 3 Oscillator Capacitor D1 1N5406 Input Diode C4 [email protected] Aux Capacitor, E-Cap D2 1N5406 Input Diode C5 [email protected] Output Capacitor, E-Cap D3 1N5406 Input Diode C6 [email protected] Feedback Filtering Capacitor D4 1N5406 Input Diode R1 [email protected] Current Sense Resistor D5 1N4937 Aux Winding Diode R2 [email protected] OCP Sensing Resistor D6 MUR460 Boost Diode R3 [email protected] Feedback Resistor D7 1N5245 Aux 15V Zener Diode R4 [email protected] Feedback Resistor Q1 MTP4N50E Power MOSFET R5 [email protected] Gate Resistor L1* 0.235mH Inductor * E 20/10/6, N67 Material from EPCOS Primary - 71 turns of # 23 AWG, Secondary - 8 turns of # 23 AWG. Gap length 0.865mm total for a primary inductance LP of 0.235mH. Figure 7. 80W universal input, follower boost variable output voltage regulation level operation power factor correction circuit 1 2 3 4 8 7 6 5 MC33260 C5 + D5 Q1R5 R4R3 R2 C1 D1 D3 D2 D4 + C4 R6 D5 D7 R7 R1 C2 C3 L1 C6 AND8016/D http://onsemi.com 10 Design Table for Universal Input, Follower Boost Variable Output Voltage Regulation Level Operation Power Factor Correction Po 25 50 75 100 125 150 200 (Watts) LP 0.752 376 0.251 0.188 0.150 0.102 0.094 (mH) Co 33 68 100 100 150 150 220 (5F) RCS 2 1 0.68 0.5 0.39 0.33 0.25 W ROCP 10000 10000 10000 9100 9100 9100 9100 W Cin 0.22 0.63 0.63 1.0 1.0 1.0 1.0 (5F) CT 0.162 0.162 0.162 0.162 0.162 0.162 0.162 (nF) Q MTD2N50E MTP4N50E MTP8N50E Dout MUR160 MUR460 Din 1N4007 1N5406 1N5406 AND8016/D http://onsemi.com 11 Notes AND8016/D http://onsemi.com 12 ON Semiconductor and are trademarks of Semiconductor Components Industries, LLC (SCILLC). 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