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Low Voltage DC Supply Dimmable Ballast for 1x36W T8 Lamp

Posted: 27 Jun 2003     Print Version  Bookmark and Share

Keywords:power 

/ARTICLES/2003JUN/A/2003JUN27_POW_AN9.PDF

Topics Covered

Introduction Schematic Diagram

Functional Description Layout Issues

Preheating & Cathode Heating Output Inductor Design

Control IC Auxilliary Component Selection Bill of Materials

Introduction

It is possible to design an effective dimmable ballast based around the IR2159 that is powered from a low voltage

DC supply instead of the AC line. A non-dimmable version based around the IR2156 is also possible using the

same basic configuration as described here. The following example shows a ballast for a single 36W T8 lamp

driven from a 30V DC supply. Lower supply voltages are possible however the IR2159 control IC can require up to

13V supplied to the VCC pin before it will operate, also as the current is high large conductors are needed to keep

losses at an acceptable level (for a 36W ballast operating from a 30V supply the input current is around 1.25A).

Functional Description

The ballast control IC and circuitry is powered from the low voltage DC supply via a simple dropper resistor that

provides the required 15.6V VCC supply from the 30V input. This can easily supply enough current for the ballast

control circuitry dissipating only a small amount of power and therefore no charge pump is needed to sustain the

supply. The circuit configuration consists of a push-pull power switching stage as opposed to the usual half-bridge

in mains supplied ballasts. This simplifies the circuit since a high side driver is not needed so the VB pin of the IC

can be connected directly to the VCC, the VS pin can be connected to 0V and no bootstrap diode or capacitor are

necessary. The output section has a step-up transformer with a split primary which produces a high voltage

switching voltage at the secondary that can be connected to a conventional inductor and capacitor ballast output

stage to the lamp. In this system both power switching MOSFET sources are connected to 0V. In order to obtain

the required current level and phase information a sense resistor must be added from the source of the LO side

MOSFET to 0V. The current will be much larger at this point than in a mains powered ballast circuit and so a lower

resistor value is necessary, in this case 0.1R. Since the step up transformer introduces no significant phase shift

and the waveform is almost identical to a current sense signal in a half bridge ballast circuit, this signal can be

used effectively to detect the phase for dimming control. In this system as well as in non-dimming designs this

point can be used to monitor the current and detect faults in the normal way allowing the ballast to shutdown if the

lamp fails to ignite correctly as in half bridge ballast circuits.

It is not necessary for the output section to be isolated from the input section and so one side of the secondary can

be referenced back to the 0V rail. We can therefore connect one side of the lamp to 0V allowing us to utilize the SD

APPLICATIONNOTE AN1038

Low Voltage DC Supply Dimmable Ballast

for 1 x 36W T8 Lamp

By Peter Green

www.irf.com 1

International Rectifier 77777 233 Kansas Street El Segundo CA 90245 USA

2 www.irf.com

AN1038

pin of the control IC to detect lamp removal or an open circuit in the lower filament. During dimming the lamp is

prone to produce striations (dark rings that appear to move along the tube). We can remove these by adding some

DC offset to the lamp voltage via R16 which is connected back to the 30V bus. A snubber consisting of R15 and

C10 is also added to reduce ringing overshoot voltages that occur when each MOSFET switches off. The snubber

will also increase the commutation time at switch off so that soft switching can be achieved using the IR2159

which has a fixed dead time of 1.8uS. The MOSFETs used are type IRF540N which have a Vdss rating of 100V

and Rds(on) of 0.044 at 25:C. The peak drain voltage is 60V plus the transient produced by the leakage induc-

tance of the step up transformer when at switch off which is comfortably less than 100V limited by the snubber.

Selection of the output L and C values

Using the lamp parameters supplied by International Rectifiers Ballast Design Assistant software we can calcu-

late the preheat, ignition and running frequencies for a bus voltage of 300V (note that version 2 and above will

calculate the frequency for 2% lamp power whereas version 1 will not). The output from the step up transformer

will be a square wave of 300Vp-p which is what would be obtained using a half bridge connected to a 300V DC

bus. The output section will be the same as in a half bridge circuit. The DC blocking capacitor must also be

included so that the step up

transformer will not drift into

saturation in one direction if the

primary volt-seconds are not

perfectly balanced in each

direction.

In this example the step up

transformer is designed to

operate at 40kHz minimum

frequency where the ballast

will be at maximum output.

The core needs to be larger to

cope with the same through-

put power at a lower frequency

so in this case in order to

limit the size to EF25 we have

chosen a 40kHz running fre-

quency. By iterating the values

of the L and C in the software

we are easily able find values

that produce the desired run-

ning frequency. The values are

L= 1.6mH and C=6.8nF.

Figure 1 The BDA software generates a curve showing the ballast operating points

www.irf.com 3

AN1038

These points are good because the preheat frequency is more than 5kHz above the ignition frequency which will

prevent the possibility of premature lamp ignition during preheat. The run frequency is as required and is well

below the ignition frequency which will allow a smooth lamp startup sequence.

Preheating and Cathode heating

Current mode pre-heating may be used in a dimming ballast designed for a 36W T8 lamp as it is able to produce

the correct preheat current which is 0.6A. The required preheat frequency can be obtained from the formula

and

where Vin will be 300V

In a dimming ballast it is also very important that the cathode current is sufficient when the lamp is dimmed.

The Cathode current at minimum can be calculated with the formula

The lamp voltage at 2% is 165Vpk and the frequency is 71kHz therefore the cathode current is 0.5Apk which is

0.35Arms.

A general rule is that the lamp filament (Cathode) resistance over the range of dimming levels should be between

3 and 5.5 times the resistance when cold. For a T8 36W lamp the cold resistance will be around 3. The resis-

tance of the cathode at maximum brightness is not critical as the arc current flowing in the lamp will serve to keep

the temperature at a sufficient level. At minimum output the cathode voltage will be around 3Vrms so the resis-

tance will be 9 which is 3 times the cold resistance. This will not be the case for many other types of lamp and

consequently voltage mode preheating would be needed.

Control IC auxiliary component selection

The quickest and easiest method for doing this in each case is to use version 2 (or higher) of the International

Rectifier Ballast Design Assistant software which can be downloaded from IR's website at www.irf.com. The BDA

supports both the IR2156 and the IR2159 as well as the IR21571 which could also be used in a non dimming low

voltage design. It can calculate the values of all external resistors and capacitors using the procedures described

below.*

(Please note that version 1 of the BDA is not suitable for using in this application)

ph

ph

ph

CV

i

f

2

=

++

-

=

22

C

ILVV

V

phinin

ph

CfVI lampcath %2%)2(

2=

4 www.irf.com

AN1038

IR2156 based system :

For non dimming design based around an IR2156 the selection of components is straightforward. CT should be

selected to provide a dead time of approximately 1.85S which is the same as the fixed dead time of the IR2159.

This can be calculated from the formula :

(Farads)

The value obtained is 1.2nF, in practise 1nF would be acceptable. The next step is to determine the value of RT

which can be calculated from the formula :

() For Frun = 40kHz the value of RT is 22K. *

The value of RPH can be calculated from the formula :

() *

The preheat time is determined by the value of CPH calculated from the formula :

(Farads) 0.33uF is the value typically used to give a 1S preheat time.

In this circuit RCS is used to shut down the circuit in a fault condition. The shutdown threshold is 1.3V therefore

the value can be calculated from the modified formula :

()

.

where Np is the transformer primary turns (center to one side) and Ns is the secondary turns.

In this case, taking a value of 2A for the ignition current the value is 0.13 so we scale this down to the nearest

preferred value 0.1.

The VDC pin can be used by connecting it via a resistor to the DC bus so that if the supply voltage falls the output

frequency will increase preventing the possibility of hard switching occurring which would cause overheating and

possible failure of the MOSFETs. A value should be chosen that will start to take effect at around 25V in a system

designed to run at 30V in this case 150K is recommended.

1475

T

T

D

C =

2892

02.1

1

-

=

runT

T

fC

R

-

-

-

=

2892

02.1

1

2892

02.1

1

PHT

T

PHT

T

PH

fC

R

fC

R

R

6

10385.0 -

W= PHPH TC

SIGN

P

CS

NI

N

R

3.1

=

www.irf.com 5

AN1038

IR2159 based system :

In a dimming system the selection of external components is more critical and the procedure more complicated.

The values of Fph, Fign, F(100%) and F(2%) can be calculated by hand using the procedure described in the

Lighting Ballast Control IC Designer's Manual 2001* based on known values of lamp voltage and power. However

the BDA software is able to do this far more quickly using lamp parameters from its own database.

* Please note that there is an error on p.213 of the Lighting Ballast Control IC Designer's Manual (2001) the

formulae for f(100%) and f% should be as stated on p250 and p.251

The following procedure can be used to determine the values for the IR2159 external resistors. Firstly Fmin must

be set which limits the minimum frequency that the oscillator will run at. This must be lower than F(100%) or Fign,

whichever is lower.

()

In this case we can select Fmin to be 40kHz this will limit the maximum ballast power. This gives a value of 36K.

The next step is to determine RCS from the formula

()

For an ignition current of 2A the value will be 0.16 which can be scaled down to 0.15 or 0.1. The over current

shutdown will obviously be more sensitive for 0.15 so if no problems of false tripping are experienced this would

give better protection.

If using the BDA software to calculate the resistor via the Program IC function it should be noted that the step up

function of the transformer will not be taken into account. Therefore the value given for RCS will need to be

multiplied by Np/Ns to provide the correct value to use in a low voltage system. This is unlikely to result in a

preferred E24 resistor value. The easiest way around this is to choose the nearest preferred value and multiply Ns/

Np. Make a note of the result and then adjust the ignition current in the software and recalculate until the RCS is the

same as the value required. Notice that the values of Riph, Rmax and Rmin will change as RCS changes. In this

example the value of 0.15 is multiplied by 125/25 giving 0.75. The ignition current is changed from 1.8A to 2.2A

and the resistor values recalculated to give RCS of 0.75.

( )

( ) 14

106

10210000

10100001025

-

--

W-

--W

=

MIN

MIN

FMIN

f

f

R

SIGN

P

CS

NI

N

R

6.1

=

6 www.irf.com

AN1038

The next step is to calculate Riph from the following formula

()

For a preheat current of 0.6Arms which is correct for this lamp the result is 22K.

In order to program the MIN and MAX settings of the dimmer interface, the phase of the output current stage at

minimum and maximum lamp power must be calculated. This is a very complicated calculation requiring the lamp

voltage and power to be known at minimum and maximum dim settings. The following method avoids the need for

this by assuming that the phase will be very close to -90: at minimum brightness and using this value to calculate

Rmin from the formula

() This give the result 27K

The BDA software can calculate the value of Rmax by first calculating the phase shift based on its database lamp

data parameters, however to get a rough estimate of what Rmax should be we can use an approximation of the

phase shift. We know from graph of Fig 1 that the frequency at 100% power is below resonance and so the phase

shift must be between 45: and zero. To obtain a starting point we can estimate a phase shift of 30: at maximum

brightness and use the formula

() This give the result 18K

Having determined the values the circuit can then be assembled and tested. By measuring the input current to the

ballast it is possible to know whether the maximum power is correct (i.e. the current is 1.25A) and if not to change

the value of Rmax accordingly. It is possible to observe the lamp brightness at minimum and adjust the value of

Rmin to obtain the desired result. It is a good idea to build the board fitting multi turn trimmers to R4 (Rmax) and R7

(Riph) to allow fine tuning of the values to give best possible results. (This can be done for Rt and Rph in the non-

dimming IR2156 based version.)

* If necessary values can be altered slightly to achieve fine adjustment of the frequencies to obtain the optimum

preheat level and lamp running power.

P

SPHCSFMIN

IPH

N

NIRR

R

2

=

-=

45

1

4

%2FMIN

MIN

R

R

--

=

45

14

86.0

%100

FMINMIN

MINFMIN

MAX

RR

RR

R

www.irf.com 7

AN1038

Schematic Diagram

Layout Issues

When laying out the PCB for this type of ballast it is important to allow the high current carrying tracks to be as wide

and as short as possible. The control IC COM should be a star point connected to the COM end of R12. The IC

COM should be connected directly to this point via a very short track. The negative side of C2 should also be

connected as close to this point as possible and the positive side should be connected as close to the center tap

of T1 as possible. The C9 decoupling capacitor should be connected directly between VCC and COM and C3, C4,

C5, R4, R5, R6, R7, C6, C7, C8 should all be connected back to the star point. Tracks around the IC should be

kept short as far as possible except the gate drives and VS and VB which can be a little longer if necessary. It is

also important to keep tracks that are carrying high switching currents away from sensitive components around

the IC as much as possible.

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

IR2159

VDC

CPH

DIM

MAX

MIN

FMIN

IPH

LO

COM

VCC

VB

VS

HO

SD

CS

VCO

1uF

C1 C2

220uF

L1

Control Input

0 to 5Vdc

(w.r.t. 0V)

+30VDC

0V

36W

T8

LAMP

R1

24K

R2

5K6

R3 10K

R4 12K

R5 27K

R6 36K

R7 27K

R8

1K

0.5W

R9

2M2

R10

680K

R11

1K

R12

0R15

R13

18R

R14

18R

Q1

IRF540N

Q2

IRF540N

R15

1K5

C3100nF

C4 10nF

C5 330nF

C6

470nF

C7

470pF

C8 C9

100uF

25V

100nF

NOTE : All capacitors are rated at 50V DC unless otherwise

stated.

C10

100nF

400V

1W

C11

100nF 400V

L2

1.6mH

C12

6n8

1500V

50V 50V

25+25 : 125

R16

22K

1W

IC1

F1

2A

T1

8 www.irf.com

AN1038

Step-Up Transformer Design

Voltage across the primary winding (from drain to drain)

The above oscilloscope traces show the voltage at the drain of each of the switching MOSFETS. The drain voltage

rises to 60V when the MOSFET switches off. This is because the primary winding is center tapped and the center

point is connected to the 30V DC bus. When one MOSFET is switched on the voltage between the center point

and the drain is 30V therefore the voltage across the entire primary winding will swing from 60V in one direction to

60V in the other direction the result being 120Vp-p.

In this design we have chosen the turns ratio of the transformer to give 300Vp-p at the secondary which can be fed

into the ballast resonant output circuit. The turns ratio required can be determined as follows :

300 / 120 = 2.5

2 x 2.5 = 5

www.irf.com 9

AN1038

Therefore the turns ratio will be 1 + 1 : 5.

The core size should be selected for a throughput power of 36W at 40kHz. We have used an EF25 (E25/13/7)

core of 3C85 or N27 material which ungapped has an Al value of 1900nH and an effective area Ae of 52mm2.

Primary Volt-Seconds = 60V x 12.5uS = 750V-uS

We have chosen 25 + 25 : 125 turns.

This gives a primary inductance of 502 x 1900nH = 4.75mH.

Therefore the magnetizing current will be 750 x 10-6 / 4.75 x 10-3 = 0.16A.

The peak flux will be

= 50 x 1900 x 10-9 x 0.16 / 52 x 10-6 = 0.29T (2900 Gauss).

This shows that the core is being pushed close to saturation in each direction but will not saturate at high

temperature (see manufacturers B-H curve for the Ferrite material).

The winding wire sizes should be chosen such that they fill the winding area. The primary should have approxi-

mately twice the diameter of the secondary as the primary RMS current will be 1.25A and the secondary RMS

current will be 0.25A.

e

PKLP

A

IAN

10 www.irf.com

AN1038

Voltage across the secondary winding

Output Inductor Design

The BDA software will design the output inductor if required. It will suggest a wire diameter for a single strand,

however a multi stranded wire that has an equivalent total cross sectional are will produce lower copper losses.

Alternatively the following procedure may be used :

1. Select the winding wire

In a dimming design because the frequency goes as high as 70kHz it is necessary to use multi stranded wire in

order to keep losses due to the skin effect to a minimum. If single stranded wire is used the inductor will run at an

increased temperature when the lamp output is low.

www.irf.com 11

AN1038

Consider the RMS running current of the lamp which can be easily estimated by dividing the maximum lamp

power by the RMS lamp voltage. The RMS lamp voltage can be approximated by dividing the peak lamp voltage by

v2 in this case 100V giving 0.36A. A current density 3A/mm2 can be used to calculate to minimum cross sectional

area of conductor that will be required. In this case the result is 0.12mm2.

The skin effect must now be considered. For Copper conductors the penetration depth at a given frequency can

be calculated by the formula

(mm)

Using the maximum frequency of 70kHz this gives the result 0.24mm therefore the strands should be less than

0.24mm diameter.

The conductor area for a wire of 0.24mm is

This gives the result 0.073mm2.

A practical solution would be to use 4 strands wire that has a diameter much smaller than 0.24mm. The area for

each strand would have to be 0.03mm2 this equates to AWG 32 which has an area of 0.046mm2 including the

insulation.

2. Select the core size

The BDA uses an iterative process which tries a range of core and gap sizes finally selecting the smallest size

that can contain the winding wire without saturating during lamp ignition. This is extremely important because if the

core does saturate the resulting current pulse will be detected at the CS pin of the IC causing the ballast to shut

down. A common design error is to fail to allow for a hot re-strike condition (i.e. when the ballast has been running

and is switched off and back on again) where the Ferrite core is already at increased temperature and the satura-

tion point of the material is reduced resulting in saturation at a lower current.

To follow the procedure of the BDA by hand is time consuming and therefore it is easier to pick an option based

on experience. For a 36W ballast a reasonable starting point would be to design an inductor based on an EF25

(E25/13/7) core with a standard gap size of 1mm made of standard power grade Ferrite (type 3C85 or N27).

For this the Al value is 63nH and Ae is 52mm2. The inductance required is 1.6mH therefore

The number of turns required is 159.

f

65

=

2

4D

A =

LA

L

N =

12 www.irf.com

AN1038

The maximum ignition current is 2A so the peak flux density will be

Which gives the result 0.39T (3900 Gauss). By looking at the manufacturers curve of B against H we can see that

the material will saturate at around 0.42T at 25:C and 0.35T at 100:C. When the ballast is cold there is no possi-

bility of saturation at ignition and during a hot re-strike situation the core is unlikely to be as hot as 100:C. Therefore

this solution is acceptable as in reality the ignition voltage of the lamp will be somewhat less than 2A if the lamp is

correctly preheated. The inductor should be built and tested under worst case conditions to ensure that the lamp

will strike. If there are problems then a larger gap or larger core will be required.

The available winding area in an EF25 bobbin is 56mm2. The winding area required is

0.046 x 4 x 159 = 29.3mm2

Allowing for gaps there will be plenty of room. It is always an advantage to increase to wire size or better still add

more strands as much as possible to minimize copper losses when the lamp is running. The BDA does this

automatically.

Bill of Materials

The following component values have been selected for a 36W T8 lamp only.

e

PKL

MAX

A

INA

B =

Item

#

Qty Manufacturer Part

Number

Description Reference

1 2 I.R. IRF540 Power MOSFET Q1,2

2 1 I.R. IR2159 Ballast Control I.C. IC1

3 1 Fuse 2A F1

4 1 Capacitor 1uF 50V 105:C

Radial Electrolytic

C1

5 1 Capacitor 220uF 50V 105:C

Radial Electrolytic

C2

6 2 Capacitor 100nF 50V C3,8

7 1 Capacitor 10nF 50V C4

8 1 Capacitor 330nF 50V C5

9 1 Capacitor 470nF 50V C6

10 1 Capacitor 470pF 50V C7

11 1 Capacitor 100uF 25V 125:C

Radial Electrolytic

C9

www.irf.com 13

AN1038

WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245 Tel: (310) 252-7105

http://www.irf.com/ Data and specifications subject to change without notice. 4/3/2002

y

12 2 Capacitor 100nF 400V

Polyester

C10,11

13 1 Capacitor 6.8nF 1500V

Polypropylene

C12

14 1 Resistor 24K 0.25W R1

15 1 Resistor 5K6 0.25W R2

16 1 Resistor 10K 0.25W R3

17 1 Resistor 12K 0.25W R4

18 2 Resistor 27K 0.25W R5,7

19 1 Resistor 36K 0.25W R6

20 1 Resistor 1K 0.5W R8

21 1 Resistor 2M2 0.25W R9

22 1 Resistor 680K 0.25W R10

23 1 Resistor 1K 0.25W R11

24 1 Resistor 0R15 0.25W R12

25 2 Resistor 18R 0.25W R13,14

26 1 Resistor 1K5 0.25W R15

27 1 Resistor 22K 1W R16

28 1 Filter Inductor L1

29 1 Inductor 1.6mH EF25 L2

30 1 Transformer 1+1:5 EF25 T1

Item

#

Qty Manufacturer Part

Number

Description Reference





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