**Amplifiers/Converters**

# Design of power factor correction circuit using Greenline compact power factor controller MC33260

**Keywords:power controller
**

/ARTICLES/2000DEC/2000DEC11_AMD_AN7.PDF |

) Semiconductor Components Industries, LLC, 1999
February, 2000 - Rev. 0
1 Publication Order Number:
AND8016/D
AND8016/D
Design of Power Factor
Correction Circuit Using
GreenlineTM Compact Power
Factor Controller MC33260
Prepared by
Ming Hian Chew
ON Semiconductor Analog Applications Engineering
Introduction
The MC33260 is an active power factor controller that
functions as a boost pre-converter which, meeting
international standard requirement in electronic ballast and
off-line power supply application. MC33260 is designed to
drive a free running frequency discontinuous mode, it can
also be synchronized and in any case, it features very
effective protections that ensure a safe and reliable
operation.
This circuit is also optimized to offer extremely compact
and cost effective PFC solutions. It does not entail the need
of auxiliary winding for zero current detection hence a
simple coil can be used instead of a transformer if the
MC33260 Vcc is drawn from the load (please refer to page
19 of the data sheet). While it requires a minimum number
of external components, the MC33260 can control the
follower boost operation that is an innovative mode
allowing a drastic size reduction of both the inductor and the
power switch. Ultimately, the solution system cost is
significantly lowered.
Also able to function in a traditional way (constant output
voltage regulation level), any intermediary solutions can be
easily implemented. This flexibility makes it ideal to
optimally cope with a wide range of applications.
This application note will discuss on the design of power
factor correction circuit with MC33260 with traditional
boost constant output voltage regulation level operation and
follower boost variable output voltage regulation level
operation. For derivation of the design equations related to
the IC please refer to MC33260 data sheet.
Figure 1. Application Schematic of MC33260
1
2
3
4
8
7
6
5
MC33260
C5
+
D5
Q1R5
R4R3
R2
C1
D1 D3
D2 D4
+
C4
R6 D5
D7
R7
R1
C2
C3
L1
C6
PFC Techniques
Many PFC techniques have been proposed, boost
topology, which can operate in continuous and
discontinuous mode, is the most popular. Typically,
continuous mode is more favorable for high power
application for having lower peak current. On the other
hand, for less than 500W application, discontinuous mode
offers smaller inductor size, minimal parts count and lowest
cost. This paper will discuss design of PFC with MC33260,
which operates in critical conduction mode.
Discontinuous Conduction Mode Operation
Critical conduction mode operation presents two major
advantages in PFC application. For critical conduction
mode, the inductor current must fall to zero before start the
next cycle. This operation results in higher efficiency and
http://onsemi.com
APPLICATION NOTE
AND8016/D
http://onsemi.com
2
eliminates boost rectifier reverse recovery loss as MOSFET
cannot turn-on until the inductor current reaches zero.
Secondly, since there are no dead-time gaps between
cycles, the ac line current is continuous thus limiting the
peak switch to twice the average input current. The
converter works right on critical conduction mode, which
results in variable frequency operation.
Inductor Waveform
V
L
+ di
dt
(1)
Equation (1) is the center of the operation of PFC boost
converter where V=Vin(t), the instantaneous voltage across
the inductor. Assuming the inductance and the on-time over
each line half-cycle are constant, di is actually the peak
current, ILpk, this is because the inductor always begins
charging at zero current.
Figure 2. Inductor Waveform
Vinpk
ILpk
Iinpk
Vin(t)
IL(t)
Iin(t)
ON
OFF
MOSFET
Design Criteria
The basic design specification concerns the following:
7 Mains Voltage Range: Vac(LL) - Vac(HL)
7 Regulated DC Output Voltage: Vo
7 Rated Output Power: Po
7 Expected Efficiency, h
PFC Power Section Design
Instantaneous Input Voltage, Vin(t)
Peak Input Voltage, Vinpk
Both Vin(t) and Vinpk are related by below equation
V
in
(t) +V
inpk
sin(t) (2)
V
inpk
+ 2 Vinrms (3)where
Instantaneous Input Current, Iin(t)
Peak Input Current, Iinpk,
Both Iin(t) and Iinpk are related by below equation
I
in
(t) +I
inpk
sin(t), (4)
I
inpk
+ 2 Iinrms (5)where
Input power of the PFC circuit, Pin can be expressed in
following equation, by substituting equation (3) and (5).
P
in
+V
inrms
I
inrms
+
V
inpk
2 @
I
inpk
2 +
V
inpk
I
inpk
2
(6)
The output power, Po is given by:
Po +VoIo +P
in (7)
PFCcircuitefficiencyisneededinthedesignequation,for
low line operation, it is typically set at 92% while 95% for
high line operation. Substituting equation (6) into equation
(7),
Po +P
in
+
V
inpk
I
inpk
2
(8)
Express the above equation in term of Iinpk,
I
inpk
+ 2Po
V
inpk
+ 2 Po
V
inrms
(9)
The average input current is equal to average inductor
current, IL(avg),
I
L(avg)
+I
in (10)
It has been understood that peak inductor current, ILpk is
exactlytwicetheaverageinductorcurrent,IL(avg)forcritical
conduction operation.
I
Lpk
+2I
L(avg)
+ 2 2 Po
V
inrms
(11)
Since ILpk is maximum at minimum required ac line
voltage, Vac(LL),
I
Lpk
+ 2 2 Po
V
ac(LL)
(12)
Switching Time
In theory, the on-time, t(on) is constant. In practice, t(on)
tends to increase at the ac line zero crossings due to the
chargeonoutputcapacitorCout.LetVac=Vac(LL)forinitial
t(on) and t(off) calculations.
On-time
By solving inductor equation (1), on-time required to
charge the inductor to the correct peak current is:
t
(on)
+I
Lpk
L
P
Vinpk
(13)
Substituting equation (3) and (12) into equation (13),
results in:
t
(on)
+ 2 2 Po
V
ac(LL)
@ L
P
2 Vac(LL)
+ 2PoL
P
V2
ac(LL)
(14)
Off-time
Theinstantaneousswitchoff-time varies with the line and
load conditions, as well as with the instantaneous line
voltage. Off-time is analyzed by solving equation (1) for the
inductor discharging where the voltage across the inductor
is Vo minus Vin.
AND8016/D
http://onsemi.com
3
t
(off)
+
I
Lpk
L
P
Vo *V
inpk
sin(t)
(15)
Multiplying nominator and denominator with
Vinpksinw(t) results in:
t
(off)
+ (16)
I
Lpk
L
P
V
inpk
sin(t)
Vo *V
inpk
sin(t)
V
inpk
sin(t)
+
Vo
2 Vinpk sin()
*1
t
(on)
where wt =
The off-time, t(off) is greatest at the peak of the ac line
voltage and approaches zero at the ac line zero crossings.
Theta () represents the angle of the ac line voltage.
The off-time is at a minimum at ac line crossings. This
equation is used to calculate t(off) as Theta approaches zero.
t
(off)min
+
I
Lpk
L
P
Vo
, +00 (17)
Switching Frequency
f + 1
t
(on)
)t
(off)
(18)
Switching frequency changes with the steady state line
and load operating conditions along with the instantaneous
input line voltage. Typically, the PFC converter is designed
to operate above the audible range after accommodating all
circuit and component tolerances. 25kHz is a good first
approximation. Higher frequency operation that can
significantly reduce the inductor size without negatively
impacting efficiency or cost should also be evaluated.
The minimum switching frequency occurs at the peak of
theaclinevoltage.Astheaclinevoltagetraversesfrompeak
to zero, t(off) approaches zero producing an increase in
switching frequency.
Inductor Value
Maximum on-time needs to be programmed into the PFC
controllertimingcircuit.Botht(on)maxandt(off)maxwillbe
individually calculated and added together to obtain the
maximumconversionperiod,ttotal.Thisisrequiredtoobtain
the inductor value.
t
(on)max
+ 2PoL
P
V 2
ac(LL)
(19)
t
(off)max
+
I
Lpk
L
P
Vo *V
inpk
, () [900 (20)
The exact inductor value can be determined by solving
equation (21) by substituting equation (19) and (20) at the
selected minimum operating frequency.
t
total
+t
(on)max
)t
(off)max (21)
Equation (21) becomes
t
total
+ 2 PoL
P
Vo
V
2
ac(LL) Vo
2 * V
ac(LL)
(22)
By rearranging in term of Lp,
Lp +
t
total
Vo
2 * V
ac(LL)
V
2
ac(LL)
2 VoPo
(23)
Equation (23) can be rewritten by substituting rearranged
equation (12) in term of 2Po.
Lp +
2 t
total
Vo
2 * V
ac(LL)
Vac(LL)
VoI
Lpk
(24)
Let the switching cycle t = 405s for universal input (85 to
265Vac)operationand205sforfixedinput(92to138Vac,or
184 to 276Vac) operation.
Inductor Design Summary
The required energy storage of the boost inductor is:
W
L
+ 1
2
L
P
I
2
Lpk
(25)
The number of turns required for a selected core size and
material is:
N
P
+
L
P
I
Lpk
106
BmaxAe
(26)
where Bmax is in Teslas and Ae is in square millimeters
(mm2)
The required air gap to achieve the correct inductance and
storage is expressed by:
lgap +4p10
*7 N
2
p Ae
L
P
mm (27)
Design of Auxiliary Winding
MC33260 does not entail an auxiliary winding for zero
current detection. Hence if DC voltage can be tapped from
the SMPS or electronic ballast connected to the output of
PFC, this step can be skipped. Then an inductor is what it
needs.
The auxiliary winding will have the low frequency
(100-120Hz) variation on its peak-rectified voltage. The
auxiliary winding capacitor must be large to minimize the
droop in the Vcc of the controller. The auxiliary turn number
can be approximated with below equation:
Naux +VauxN
P
VL
+
VauxN
P
Vo *V
ac(LL) (28)
Selection of Output Capacitor
The choice of output capacitance value is dictated by the
required hold-up time, thold or the acceptable output ripple
AND8016/D
http://onsemi.com
4
voltage,Voripforagivenapplication.Asaruleofthumb,can
start with 15F/watt.
Selection of Semiconductors
Maximum currents and voltages must first be determined
for over all operating conditions to select the MOSFET and
boost rectifier. As a rule of thumb, derate all semiconductors
to about 75-80% of their maximum ratings. This implying
the need of devices with at least 500V breakdown voltage.
BipolartransistorsareanacceptablealternativetoMOSFET
if the switching frequency is maintained fairly low. High
voltage diodes with recovery times of 200ns, or less should
be used for the boost rectifier. One series of the popular
devices is the MURXXX Ultrafast Rectifier Series from ON
Semiconductor.
Maximum power MOSFET conduction losses.
P
(on)max
[1
6
R
ds(on)
I
2
Lpk
1 *
1.2 V
ac(LL)
Vo
(29)
Designing the Oscillator Circuit
For traditional boost operation, CT is chosen with below
equation:
C
T
w2 Kosc L
P
P
in
V2
o
V
2
ac(LL) R
2
o
*Cint (30)
Design of Regulation and Overvoltage
Protection Circuit
The output voltage regulation level can be adjusted by Ro,
Ro [ Vo
200 5A
(31)
Designing the Current Sense Circuit
The inductor current is converted into a voltage by
insertinga ground referenced resistor, RCS in series with the
input diode bridge. Therefore a negative voltage
proportional to the inductor current is built.
The current sense resistor losses, PRcs:
P
Rcs
+1
6
R
CS
I
2
Lpk
(32)
Overcurrent protection resistor, ROCP can be determined
with below equation:
R
OCP
+
R
CS
I
Lpk
I
OCP
(33)
Current Limiting With Boost Topology Power
Factor Correction Circuit
Unlike buck and flyback circuits, because there is no
seriesswitchbetweeninputandoutputintheboosttopology,
high current occurring with the start-up inrush current surge
charging the bulk capacitor and fault load conditions cannot
be limited or controlled without additional circuitry.
The MC33260 Zero Current Detection uses the current
sensing information to prevent any power switch turn on as
long as some current flows through the inductor. Then,
during start-up, the power MOSFET is not allowed to turn
on while in-rush current flows. Then there is no risk to have
the power switch destroyed at start-up because of the
in-rush current.
In the same way, in an overload case, the power MOSFET
is kept off as long as there is a direct output capacitor charge
current, i.e., when the input voltage is higher than the output
voltage.Consequently, overload working is fully safe for the
power MOSFET. This is one of the major advantages
compared to MC33262 and competition.
Current Limiting for Start-up Inrush
Initially Vo is zero, when the converter is turned on, the
bulk capacitor will charge resonantly to twice Vin. The
voltage can be as high as 750V if Vin happens to be at the
peak high-line 265V condition (375V). The peak resonant
charging current through the inductor will be many times
greater than normal full load current. the inductor must be
designed to be much larger and more expensive to avoid
saturation. The boost shunt switch cannot do anything to
prevent this and could be worse if turned on during start-up.
The inrush current and voltage overshoot during the
start-up phase is intolerable. A fuse is not suitable, as it will
blow each time the supply is turned on.
There are several methods that may be used to solve the
start-up problem:
1. Start-up Bypass Rectifier
This is implemented by adding an additional rectifier
bypassingthe boost inductor. The bypass rectifier will divert
the start-up inrush current away from the boost inductor as
shown in Figure 3. The bulk capacitor charges through
Dbypass to the peak AC line voltage without resonant
overshootandwithoutexcessiveinductorcurrent.Dbypassis
reverse-biased under normal operating conditions. If load
overcurrent pulls down Vo, Dbypass conducts, but this is
probably preferable to having the high current flowing
through boost inductor.
Figure 3. Rectifier bypass of start-up inrush current
VAC
VOUT
Dbypass
+PFC
IC
2. External Inrush Current Limiting Circuit
For low power system, a thermistor in series with the
pre-converter input will limit the inrush current. Concern is
the thermistor may not respond fast enough to provide
protection after a line dropout of a few cycles.
AND8016/D
http://onsemi.com
5
A series input resistor shunted by a Triac or SCR is a more
efficient approach. A control circuit is necessary. This
method can function on a cycle-by-cycle basis for
protection after a dropout.
Load Overcurrent Limiting
If an overcurrent condition occurs and exceeds the boost
converter power limit established by the control circuit, Vo
will eventually be dragged down below the peak value of the
AC line voltage. If this happens, current will rise rapidly and
without limit through the series inductor and rectifier. This
may result in saturation of the inductor and components will
fail. The control circuit holds off the shunt switch, since the
current limit function is activated. It cannot help to turn the
switch ON - the inductor current will rise even more rapidly
and switch failure will occur.
Typically, a power factor correction circuit is connected to
another systems like switched mode power supply or
electronic ballast. These downstream converters typically
will have current limiting capability, eliminating concern
about load faults. However, a downstream converter or the
bulk capacitor might fail. Hence there is a possibility of a
short circuit at the load.
If it is considered necessary to limit the current to a safe
value in the event of a downstream fault, some means
external to the boost converter must be provided.
Design Example I - Traditional Boost Constant
Output Voltage Regulation Level Operation
Power Factor Correction
The basic design specification concerns the following:
7 Mains Voltage Range: Vac(LL) - Vac(HL) = 85 - 265Vac
7 Regulated DC Output Voltage: Vo = 400Vdc
7 Rated Output Power: Po = 80W
7 Expected Efficiency, h > 90%
A. The input power, Pin is given by
P
in
+Po
+ 80
0.92
+86.96W
B. Input diode current is maximum at Vinrms =
Vac(LL)
I
inpk
+ 2 Po
V
ac(LL)
+ 2 80
0.92 85
+1.447A
C. Inductor design
1. Inductor peak current:
I
Lpk
+2I
inpk
+2 1.447 +2.894A
2. Inductor value:
Lp +
2 t
total
Vo
2 * V
ac(LL)
Vac(LL)
VoI
Lpk
+
2 40 10*6 400
2 * 85 85
400 2.894
+1.162mH
Let the switching cycle t = 40ms for universal input (85 to
265Vac) operation.
3. The number of turns required for a selected core size
and material is:
N
P
+
L
P
I
Lpk
106
BmaxAe
+1.162 10*3 2.894 10*6
0.3 60
+186.8 turns [187 turns
Using EPCOS E 30/15/7, Bmax =0.3T and Ae = 60mm2.
4. The required air gap to achieve the correct inductance
and storage is:
lgap +4p10
*7
N
2
pAe
L
P
+4p 10*7 1872 60 10*6
1.162 10*3
+2.269mm
5. Design of Auxiliary Winding
Naux + VauxN
P
Vo *V
ac(LL)
+ 14 187
(400 *265)
+19.4 turns [20 turns
Round up to 20 turns to make sure enough voltage at the
auxiliary winding.
D. To determine the output capacitor
As rule of thumb, for 80W output, start with 100mF, 450V
capacitor.
E. Calculation of MOSFET conduction losses
A 8A, 500V MOSFET, MTP8N50E is chosen. The on
resistance, Rds(on)[1.75W@1000C. Therefore, maximum
power MOSFET conduction losses is:
P
(on)max
[1
6
R
ds(on)
I
2
Lpk 1 *
1.2 V
ac(LL)
Vo
+1
6
1.75 2.8942 1 *1.2 85
400
+1.82W
F. Design of regulation and overvoltage
protection circuit
The output voltage regulation level can be adjusted by Ro,
Ro [ Vo
200 5A
+ 400
200 5A
+2M
Use two 1MW resistors in series.
G. Designing the oscillator circuit
For traditional boost operation, CT is chosen with below
equation:
C
T
w2 Kosc L
P
P
in
V2
o
V2
ac(LL) R2
o
*Cint +
2 6400 1.162mH 86.96 4002
852 2M2
*15pF +7.16nF
AND8016/D
http://onsemi.com
6
Use 10nF capacitor.
Figure 4. Theoretical Vo versus Vac with CT = 10nF
Vo/(V)
85 115 175
85
145
Vac (V)
145
175
115
445
205
235
205 235
265
295
325
355
385
415
265100 1130 190160 220 250 280
Full Load
Half Load
Vacpeak
H. Design of the current sense circuit
Choose Rcs = 0.68W
1. So the current sense resistor losses, PRcs:
P
Rcs
+1
6
R
CS
I
2
Lpk +1
6
1 2.894
2
+0.949W
Therefore the power rating of RCS is chosen to be 2W.
2. Overcurrent protection resistor, ROCP can be
determined with below equation:
R
OCP
+
R
CS
I
Lpk
I
OCP
+0.68 2.894
205 5A
+9600
Use 10000W resistor. This provide current limit at 3.01A
versus calculated value of ILpk = 2.894A.
80W, Universal Input, Traditional Boost Constant Output Voltage Level Regulation Operation Power Factor
Correction Circuit Part List
Index Value Comment Index Value Comment
C1 0.635F@600V Filtering Capacitor R6 22W@0.25W Aux Winding Resistor
C2 680nF Pin 2 Vcontrol Capacitor R7 100KW@2W Start-up Resistor
C3 10nF Pin 3 Oscillator Capacitor R8 1N5406 Input Diode
C4 1005F@50V Aux Capacitor, E-Cap D1 1N5406 Input Diode
C5 1005F@450V Output Capacitor, E-Cap D2 1N5406 Input Diode
C6 1nF@50V Feedback Filtering Capacitor D3 1N5406 Input Diode
R1 0.68W@2W Current Sense Resistor D4 1N4937 Aux Winding Diode
R2 10KW@0.25W OCP Sensing Resistor D5 MUR460 Boost Diode
R3 1MW@0.25W Feedback Resistor D6 1N5245 Aux 15V Zener Diode
R4 1MW@0.25W Feedback Resistor D7 MTP8N50E Power MOSFET
R5 10W@0.25W Gate Resistor Q1 1.162mH Inductor
* E 30/15/7, N67 Material from EPCOS
Primary - 187 turns of # 23 AWG, Secondary - 19 turns of # 23 AWG.
Gap length 2.269mm total for a primary inductance LP of 1.162mH.
AND8016/D
http://onsemi.com
7
Figure 5. 80W universal input, traditional boost constant output voltage regulation level operation power factor
correction circuit
1
2
3
4
8
7
6
5
MC33260
C5
+
D5
Q1R5
R4R3
R2
C1
D1 D3
D2 D4
+
C4
R6 D5
D7
R7
R1
C2
C3
L1
C6
Design Table for Universal Input, Traditional Boost Constant Output Voltage Regulation Level Operation Power
Factor Correction
Po 25 50 75 100 125 150 200 (Watts)
LP 3.720 1.860 1.240 0.930 0.744 0.620 0.465 (mH)
Co 33 68 100 100 150 150 220 (5F)
RCS 2 1 0.68 0.5 0.39 0.33 0.25 W
ROCP 10000 10000 10000 9100 9100 9100 9100 W
Cin 0.22 0.63 0.63 1.0 1.0 1.0 1.0 (5F)
CT 10 10 10 10 10 10 10 (nF)
Q MTP4N50E MTP8N50E MTW14N50E
Dout MUR160 MUR460
Din 1N4007 1N5406
Design Example II - Follower Boost Variable
Output Voltage Regulation Level Operation
Power Factor Correction
The basic design specification concerns the following:
7 Mains Voltage Range: Vac(LL) - Vac(HL) = 85 - 265Vac
7 Maximum Regulated DC Output Voltage: Vo = 400Vdc
7 Minimum Regulated DC Output Voltage: Vomin =
140Vdc
7 Rated Output Power: Po = 80W
7 Expected Efficiency, h > 90%
A. The input power, Pin is given by
P
in
+Po
+ 80
0.92
+86.96W
B. Input diode current is maximum at Vinrms =
Vac(LL)
I
inpk
+ 2 Po
V
ac(LL)
+ 2 80
0.92 85
+1.447A
C. Inductor design
1. Inductor peak current:
I
Lpk
+2I
inpk
+2 1.447 +2.894A
2. Inductor value, for follower boost operation, Vo =
Vomin:
Lp +
2 t
total
Vomin
2 * V
ac(LL)
V
omin
I
Lpk
+
2 40 10*6 140
2 * 85 85
140 2.894
+0.235 5H
Let the switching cycle t = 405s for universal input (85 to
265Vac) operation.
3. The number of turns required for a selected core size
and material is:
AND8016/D
http://onsemi.com
8
N
P
+
L
P
I
Lpk
106
BmaxAe
+
0.235 10*3 2.894 106
0.3 32.1
+70.6 turns [71 turns
Using EPCOS E 20/10/6, N67 material, Bmax =0.3T and
Ae = 32.1mm2.
4. The required air gap to achieve the correct inductance
and storage is:
lgap +4p10
*7
N
2
pAe
L
P
+4p 10*7 712 32.1 10*6
0.235 10*3
+0.856mm
5. Design of Auxiliary Winding
Naux + VauxN
P
Vo *V
ac(LL)
+ 14 71
(400 *265)
+7.4 turns [8 turns
Round up to 8 turns to make sure enough voltage at the
auxiliary winding.
D. To determine the output capacitor
As rule of thumb, for 80W output, start with 1005F, 450V
capacitor.
E. Calculation of MOSFET conduction losses
A 4A, 500V MOSFET, MTP4N50E is chosen. The on
resistance, Rds(on)[1.75W@1000C. Therefore, maximum
power MOSFET conduction losses is:
P
(on)max
[1
6
R
ds(on)
I
2
Lpk 1 *
1.2 V
ac(LL)
V
omin
+1
6
1.75 2.8942 1 *1.2 85
140
+0.66W
F. Design of regulation and overvoltage
protection circuit
The output voltage regulation level can be adjusted by Ro,
Ro [ Vo
200 5A
+ 400
200 5A
+2M
Use two 1MW resistors in series.
G. Designing the Oscillator Circuit
For follower boost operation, CT is chosen with below
equation:
C
T
w2 Kosc L
P
P
in
V2
o
V2
ac(LL) R2
o
*Cint +
2 6400 0.234mH 86.96 1402
852 2M2
*15pF +162pF
Use 150pF capacitor.
Figure 6. Theoretical Vo versus Vac with CT = 150pF
Vo/(V)
85 115 175
85
145
Vac (V)
145
175
115
445
205
235
205 235
265
295
325
355
385
415
265100 1130 190160 220 250 280
Full Load
Half Load
Vacpeak
H. Design of the Current Sense Circuit
Choose Rcs = 0.68W
1. So the current sense resistor losses, PRcs:
P
Rcs
+1
6
R
CS
I
2
Lpk
+1
6
0.68 2.8942 +0.949W
2. Overcurrent protection resistor, ROCP can be
determined with below equation:
R
OCP
+
R
CS
I
Lpk
I
OCP
+0.68 2.894
205 5A
+9600
Use 10000W resistor. This provide current limit at 3.01A
versus calculated value of ILpk = 2.894A.
AND8016/D
http://onsemi.com
9
80W, Universal Input, Follower Boost Variable Output Voltage Regulation Level Operation Power Factor
Correction Circuit Part List
Index Value Comment Index Value Comment
C1 0.635F@600V Filtering Capacitor R6 22W@0.25W Aux Winding Resistor
C2 680nF Pin 2 Vcontrol Capacitor R7 100KW@2W Start-up Resistor
C3 150pF Pin 3 Oscillator Capacitor D1 1N5406 Input Diode
C4 1005F@50V Aux Capacitor, E-Cap D2 1N5406 Input Diode
C5 1005F@450V Output Capacitor, E-Cap D3 1N5406 Input Diode
C6 1nF@50V Feedback Filtering Capacitor D4 1N5406 Input Diode
R1 0.68W@2W Current Sense Resistor D5 1N4937 Aux Winding Diode
R2 10KW@0.25W OCP Sensing Resistor D6 MUR460 Boost Diode
R3 1MW@0.25W Feedback Resistor D7 1N5245 Aux 15V Zener Diode
R4 1MW@0.25W Feedback Resistor Q1 MTP4N50E Power MOSFET
R5 10W@0.25W Gate Resistor L1* 0.235mH Inductor
* E 20/10/6, N67 Material from EPCOS
Primary - 71 turns of # 23 AWG, Secondary - 8 turns of # 23 AWG.
Gap length 0.865mm total for a primary inductance LP of 0.235mH.
Figure 7. 80W universal input, follower boost variable output voltage regulation level operation power factor
correction circuit
1
2
3
4
8
7
6
5
MC33260
C5
+
D5
Q1R5
R4R3
R2
C1
D1 D3
D2 D4
+
C4
R6 D5
D7
R7
R1
C2
C3
L1
C6
AND8016/D
http://onsemi.com
10
Design Table for Universal Input, Follower Boost Variable Output Voltage Regulation Level Operation Power
Factor Correction
Po 25 50 75 100 125 150 200 (Watts)
LP 0.752 376 0.251 0.188 0.150 0.102 0.094 (mH)
Co 33 68 100 100 150 150 220 (5F)
RCS 2 1 0.68 0.5 0.39 0.33 0.25 W
ROCP 10000 10000 10000 9100 9100 9100 9100 W
Cin 0.22 0.63 0.63 1.0 1.0 1.0 1.0 (5F)
CT 0.162 0.162 0.162 0.162 0.162 0.162 0.162 (nF)
Q MTD2N50E MTP4N50E MTP8N50E
Dout MUR160 MUR460
Din 1N4007 1N5406 1N5406
AND8016/D
http://onsemi.com
11
Notes
AND8016/D
http://onsemi.com
12
ON Semiconductor and are trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes
withoutfurther notice to any products herein. SCILLC makes no warranty,representationorguaranteeregardingthesuitabilityofitsproductsforanyparticular
purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability,
including without limitation special, consequential or incidental damages. "Typical" parameters which may be provided in SCILLC data sheets and/or
specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including "Typicals" must be
validated for each customer application by customer's technical experts. SCILLC does not convey any license under its patent rights nor the rights of others.
SCILLCproducts are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications
intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or
death may occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold
SCILLC and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable
attorneyfees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim
alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal Opportunity/Affirmative Action Employer.
PUBLICATION ORDERING INFORMATION
ASIA/PACIFIC: LDC for ON Semiconductor -- Asia Support
Phone: 303-675-2121 (Tue-Fri 9:00am to 1:00pm, Hong Kong Time)
Toll Free from Hong Kong 800-4422-3781
Email: ONlit-asia@hibbertco.com
JAPAN: ON Semiconductor, Japan Customer Focus Center
4-32-1 Nishi-Gotanda, Shinagawa-ku, Tokyo, Japan 141-8549
Phone: 81-3-5487-8345
Email: r14153@onsemi.com
Fax Response Line: 303-675-2167
800-344-3810 Toll Free USA/Canada
ON Semiconductor Website: http://onsemi.com
For additional information, please contact your local Sales Representative.
AND8016/D
Greenline is a trademark of Semiconductor Components Industries, LLC.
North America Literature Fulfillment:
Literature Distribution Center for ON Semiconductor
P.O. Box 5163, Denver, Colorado 80217 USA
Phone: 303-675-2175 or 800-344-3860 Toll Free USA/Canada
Fax: 303-675-2176 or 800-344-3867 Toll Free USA/Canada
Email: ONlit@hibbertco.com
N. American Technical Support: 800-282-9855 Toll Free USA/Canada
EUROPE: LDC for ON Semiconductor -- European Support
German Phone: 303-308-7140 (Mon-Fri 2:30pm to 5:00pm Munich Time)
German Email: ONlit-german@hibbertco.com
French Phone: 303-308-7141 (Mon-Fri 2:30pm to 5:00pm Toulouse Time)
French Email: ONlit-french@hibbertco.com
English Phone: 303-308-7142 (Mon-Fri 1:30pm to 5:00pm UK Time)
English Email: ONlit@hibbertco.com

Related Articles | Editor's Choice |

Visit Asia Webinars to learn about the latest in technology and get practical design tips.